band scopes). This should handle the normal range of signal
levels encountered in high frequency circuitry. It will not
handle very low amplitude signals applicable to radio
receiver circuits, but that is a job intended for spectrum
analyzers only.
As to circuit loading, there really are no high impedance
circuits once we enter the VHF spectrum and above.
Normally, most circuit source impedances will range from
25-500 ohms at these frequencies. Even reactive LC circuitry
will have much lower impedance comparable to low
frequency counterparts. So, the 3,400 ohm loading factor of
this probe will pose an acceptably small loading on most
VHF circuitry. And more importantly, it imposes a very small
reactive loading (>>1 pF). In high frequency measurement,
the probe’s reactive component is much more critical than
its resistive one in terms of circuit loading (as explained
earlier). Finally, as to external power, this turned out to be of
little inconvenience due to the amplifier’s onboard regulator
and power plug. It required only a separate wall plug
transformer of adequate voltage. I leave this wall wart
plugged into an outlet strip mounted on the back of my
scope cart. This makes probe installation quick and easy —
just one added cable to connect/disconnect.
Probe Design
In designing this probe, a number of factors had to be
considered. It all starts with the MMIC (monolithic
microwave integrated circuit) amplifier which has a gain of
31 db (X34), a 1 db compression point (the very onset of
saturation) of 12. 5 dbm (approx. 2.5V p-p), and a frequency
response roll-off of - 4 db at 500 MHz. It also requires a 50
ohm source and load impedance to maintain these specs,
along with good stability. Best performance with these amps
is obtained with a bias (B+) current of 36 mA. The B+
supplying this current ideally would be through a load
resistance of infinite impedance because this load is
effectively placed in parallel with the 50 ohm termination
impedance. A constant current source comes to mind but
will not work here as per manufacturer’s application notes.
If the bandwidth ratio was not quite so wide, a suitable RFC
(radio frequency choke) could be used with a bias supply of
5 VDC. However, this circuit requires a very wide
bandwidth ratio (1,000:1); no RFC would work due to their
inherent self-resonance at some point, rendering them
useless at frequencies above that. So, a pure resistance is
ITEM
C1
C2
C3
C4
C5
C6
R1
R2
R3
R4
MMIC
18V Regulator
Wall plug
Power Pack
required here to supply the bias current and I want that
value to be as high as possible. Of course, the higher the
resistance, the higher the supply voltage has to be to supply
that current. A compromise had to be made here in terms
of voltage, wattage dissipation, and shunt loading. I chose
18 volts which works out to 392 ohms for the supply
resistor with a dissipation of 1/2 watt, without upsetting the
output termination to a great degree.
Given that 392 ohms is not the ideal value here, the
amp’s maximum output level is degraded, so I decided to
spec that figure at + 10 dbm ( 2.0V p-p) for best linearity.
Given the amplifier’s gain with a 1:1 attenuation ratio, this
would only allow for a max probe input impedance of 850
ohms and its dynamic range limited to 2.0V p-p on the
upper end. These limits just would not satisfy a wide range
of probing situations, but by increasing probe attenuation to
4:1, I could get the 8V p-p input level I was after at the
upper end of range, and increase the input impedance to
3,400 ohms. The actual input impedance works out to
3,400 ohms shunted by <<1 pF. This provides an acceptable
loading factor while maintaining the desired dynamic range.
It now becomes a 4x probe and it is easy to calculate
the actual display amplitude vs. input signal amplitude — just
double the displayed voltage and double it again (X4). Not
as easy as a 1X or 10X probe, but still quite simple to
compute in one’s head. The amplifier roll off, cabling
connections, and board layout combined will add up to a
little less than a - 6 db (2x) loss at the upper limit of 500
MHz of the probe’s bandwidth, so the next problem to be
solved was how to flatten this response curve. The
straightforward answer was to shunt R1 and R2 with a
compensation capacitance. The Xc needed for this
compensation worked out to be 2,000 ohms at 500 MHz
which has a value of 0.18 pF. It worked out in my favor on
this one due to the fact that 1/4 watt carbon film resistors
have a parasitic capacitance of 0.35 pF. Two in series would
give me 0.175 pF — almost exactly what I needed! A call to
KOA Speer’s (resistor manufacturer) engineering department
confirmed that these resistors are made to exacting
mechanical tolerances and that the parasitic is uniform from
unit to unit with one minor exception. The laser etching
used in the manufacturing process is slightly different from
one group of values to the next; a group value being a
certain range of resistance in perhaps a 5:1 ratio.
Next I needed to pick out two values of resistance that
would give me a combined value of 3,400 ohms and a
parasitic capacitance of
approximately 0.175 pF. The
values chosen were
accomplished empirically
due to the fact that the
exact capacitance could not
be predetermined. After
considerable
experimentation, a set of
values was found to be
1,000 ohms in series with
2,400 ohms, with the
correct parasitic capacitance
to flatten out the response
DESCRIPTION SOURCE
O.1 MFD/50V MLC ceramic
0.22 MFD/50V chip size 1812
0.47 MFD/50V chip size 1812
0.33 MFD/50V MLC ceramic
1,000 pF/50V MLC ceramic
4. 7 MFD/50V tantalum electrolytic
1,000 ohm 1/4W carbon film Mouser.com p/n 600 - CF 1/4 102J
2,400 ohm 1/4W carbon film Mouser p/n 600 - CF 1/4 242J
51 ohm 1/4W carbon film
392 ohm 1W 1% metal film
MAR-8A mini-circuits
78L18
Active Probe
Parts List
Mouser p/n 71 - CMF60392R00FHEK
Triad 18V/300 mA
Mouser p/n 553-WDU18-300
48
November 2010